Voltage regulated power supply utilizing phase shift control

ABSTRACT

A power supply, including a resonant circuit having an output voltage and a current oscillating therethrough, and a voltage-fed half-bridge inverter producing a source voltage at an output coupled to the resonant circuit. The inverter is responsive to a driving signal. A driving circuit has a first input representing the sensed current oscillating through the resonant circuit, a second input representing the output voltage, and a reference voltage. The driving circuit includes compensation circuitry for maintaining the output voltage at the reference voltage and commanding a phase shift angle, and phase-shifting circuitry producing the driving signal based on a phase-shift of the sensed current. The amount of phase shift is commanded by the compensation circuitry.

CROSS REFERENCE TO RELATED APPLICATIONS

This is a continuation-in-part of U.S. application Ser. No. 08/773,243,filed Dec. 23, 1996 now U.S. Pat. No. 5,781,418, which is incorporatedby reference herein.

BACKGROUND OF THE INVENTION

1. Technical Field

The invention generally concerns power supplies and lighting ballasts,and more particularly to voltage regulation for zero-voltage-switched(ZVS) inverters.

2. Description of the Prior Art

Electronic lamp ballasts include inverters to provide a high frequencysubstantially square wave voltage output to a load. The load typicallyincludes a resonant circuit and lamps. A typical inverter circuitincludes power switching circuitry involving transistors to convert a DCinput to a desired high frequency AC output.

A control circuit, such as disclosed in U.S. Pat. No. 4,952,849 toFellows et al., for driving the switching circuitry of a voltage-fedinverter, senses current flowing through the load to control theswitching circuitry. The switches are turned on so as to minimize damageand losses thereto during transition and is commonly referred to aszero-voltage switching (ZVS). The switching technique minimizes thevoltage across the switch as the switch is being turned on.

The switching frequency of the inverter is typically above the resonantfrequency of the resonant circuit, that is, to maintain the resonantcircuit in an inductive mode. More generally, in the zero-voltageswitching scheme, it is critical to maintain the circuit in an inductivemode when transitioning between the power switches, regardless of theload type. Otherwise, large power losses and damage to the componentswithin the circuit can result.

One commonly used method to achieve zero-voltage switching is tomaintain the switching frequency higher than the resonant frequency, asdescribed in Steigerwald, "A Comparison of Half-Bridge ResonantConverter Topologies," IEEE Transactions on Power Electronics, April1988, pp. 174-181, incorporated by reference herein. Thisfrequency-based scheme can be easily implemented using voltagecontrolled oscillators (VCO). However, a critical requirement of such acontrol method is that prior knowledge of the resonant frequency isneeded to determine the frequency sweeping region. In simpleapplications, for example, in LC resonant inverter-based power suppliesand electronic ballasts with a single lamp, when the maximum gainfrequency does not change significantly during the load and linechanges, the frequency-based control method operates successfully.

However, in some cases, the characteristics of the resonant circuit canchange rapidly with lamp loads, bus voltage, ambient conditions or agingof the components. Such changes can affect the operating mode of theresonant circuit. Simple frequency based control methods may be unableto adapt to the new frequency-sweep region and therefore be unable tomaintain zero voltage switching.

The circuit of Nalbant, "A New and Improved Control Technique GreatlySimplifies the Design of ZVS Resonant Inverters and DC/DC PowerSupplies," Proceedings of IEEE Applied Power Electronics Conference,March 1995, pp. 694-701, achieves zero voltage switching by increasingthe operating frequency above the resonant frequency. Switching occursby setting thresholds on the current flowing in a loading matchingnetwork. When one of these thresholds is crossed, switching occurs inthe inverter. This approach has the disadvantage that threshold valuesmust be chosen and set, and complex logic must be implemented. If thecurrent values experience large fluctuations, the threshold may not becrossed and a correct switching cycle may be missed, or the wrongoperating frequency may result. Here, the inductor current informationis not further used for output regulation.

Accordingly, it is an object of the invention to provide various voltageregulation methods in a zero-voltage switching scheme for a power supplyor ballast having a voltage-fed inverter which overcomes the abovementioned disadvantages of the prior art.

SUMMARY OF THE INVENTION

A high frequency resonant power supply is disclosed which includes aresonant circuit having an output voltage and a current oscillatingtherethrough, a voltage-fed half-bridge inverter producing a sourcevoltage at an output coupled to the resonant circuit, the inverter beingresponsive to a driving signal. The power supply further includes anovel driving circuit producing the driving signal.

The driving circuit has compensation circuitry regulating the outputvoltage with respect to a reference voltage and commanding a phase shiftangle. Phase-shifting circuitry in the driving circuit applies aphase-shift to the sensed current, and the amount of phase shift iscommanded by the compensation circuitry.

The phase-shifting circuitry applies the positive phase angle shift tothe sensed current input such that the voltage at the output of theinverter is leading the sensed current input with the predeterminedpositive phase angle. The driving signal may actuate a switch in theinverter when the phase-shifted sensed current signal crosses zero. Thephase shifting circuitry preferably includes an all-pass filter.

In a preferred embodiment, the driving circuit includes a filter forsuppressing a low frequency component of the sensed current flowingthrough the resonant circuit. The power supply may further include aload, wherein the load is a discharge lamp.

In a preferred embodiment, the driving circuit includes a hysteresiscontrol with respect to the sensed current signal. The hysteresiscontrol is used to establish the switching times. The hysteresisrelation is so designed that the zero-voltage switching is alwaysachieved at each switching time, regardless of the load situation.

A method for switching a power supply is also disclosed which includesthe steps of providing a resonant circuit having a current oscillatingtherethrough; providing an inverter responsive to a driving signal andproducing a voltage at an output coupled to the resonant circuit;sensing the current oscillating through the resonant circuit; applying aphase angle shift to the current; applying a driving signal to theinverter based upon the phase-shifted current.

In a preferred embodiment, the method further includes the step offiltering the sensed current signal to suppress low frequency componentsprior to applying the phase angle shift. In the method for switching apower supply, the step of applying a driving signal to the inverterpreferably includes actuating a switch in the inverter when thephase-shifted current crosses zero.

A method for switching a power supply is also disclosed which includesthe steps of providing a resonant circuit having a current oscillatingtherethrough; providing an inverter responsive to a driving signal andproducing a voltage at an output coupled to the resonant circuit;sensing the current oscillating through the resonant circuit andproducing a sensed current signal proportional to the current; applyinga phase angle shift to the sensed current signal and producing aphase-shifted signal; and applying a driving signal to the inverterbased upon zero-crossing points of the phase-shifted signal.

It is an object of the invention to provide a voltage regulation methodusing a phase shift of the signal, for example, for the signal appliedto the resonant inverters.

It is an object of the invention to provide a voltage regulation methodusing a level shifted control reference compared with the sensed currentsignal, for example, for the signal applied to the resonant inverters.

It is a further object of the invention to provide a voltage regulationmethod, using a level-shifted control reference multiplied with thesensed current signal, for example, for the signal applied to theresonant inverters.

It is an object of the invention to produce an improved zero-voltageswitching scheme which employs a phase-shifting technique in timing theswitching cycles in the inverter.

It is another object of the invention to provide a switching schemewhich does not require prior identification of the resonant frequency.

It is a further object of the invention to provide a general switchingscheme which is applicable to resonant circuits of second order andhigher.

It is yet another object of the invention to provide a switching schemewhich does not require complex threshold logic for the timing of theswitches.

These and other objects, features and advantages of the invention willbecome apparent with reference to the accompanying drawings and thefollowing detailed description and claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of the power supply in accordance with anexemplary embodiment of the subject invention;

FIG. 2(a) is a curve illustrating normalized frequency Ω(φ) as afunction of the phase angle φ between current and voltage;

FIG. 2(b) is a curve illustrating circuit voltage gain as a function ofthe phase angle between current and voltage;

FIG. 3 is time plot of the current flowing through the resonant circuitand voltage;

FIG. 4(a) is a plot of the relationship between voltage gain andfrequency for a lamp load;

FIG. 4(b) is a plot of the relationship between phase angle andfrequency for a lamp load;

FIG. 5 is a plot of the relationship between voltage gain and phaseangle for a lamp load;

FIG. 6 is plot of the hysteresis function to achieve the switchingcycling of FIG. 3;

FIG. 7 is a schematic diagram of the inverter and the power supply;

FIG. 8 is a block diagram of the power supply, illustrating theparticular functions of the driving circuit;

FIG. 9 is a schematic diagram of a portion of the driving circuit,illustrating the current signal amplifier, the high-pass filter, gaincontrol, phase-shift circuitry;

FIG. 9A is a schematic diagram of the voltage feedback and regulationcircuitry;

FIG. 10 is a partial schematic, partial logic diagram of a portion ofthe driving circuit, illustrating the wave-shaping and PWM functions;

FIG. 10A is partial block diagram, partial schematic diagram of thetimer circuit of FIG. 10;

FIG. 11 is a schematic diagram illustrating the start-up circuitry;

FIG. 12 is a schematic diagram illustrating the initiation frequencysweep function associated with start-up;

FIG. 13 is a schematic diagram illustrating the bridge driver circuitry;

FIG. 13A is a schematic diagram illustrating the functions of the driverIC;

FIG. 14 is a time plot of the current flowing through the resonantcircuit, the phase-shifted signal, and the switching sequence of theswitches;

FIG. 15 is a schematic diagram of an alternative embodiment of thesubject invention;

FIG. 16 is a schematic diagram of a second alternative embodiment of thesubject invention; and

FIG. 17 is a schematic diagram of a third alternative embodiment of thesubject invention;

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The power supply circuit shown in FIG. 1 includes a voltage source V,inverter circuit A, resonant circuit and load B, and driving circuit Cwhich controls inverter circuit A. Voltage source V is a direct currentvoltage source supplying inverter circuit A via lines RL1 and RL2.Alternatively, an AC voltage source may be substituted in conjunctionwith a rectifier circuit, which may further include power factorcorrection. Inverter circuit A is of the voltage-fed half-bridgetopology, and is under control of the driving circuit C. Resonantcircuit B receives the substantially square wave output of invertercircuit A on lines 100 and 102 and converts it into a substantiallysinusoidal lamp current. In the exemplary embodiment, the load is aparallel-loaded lamp system.

The driving circuit C receives a voltage signal proportional to thecurrent flowing through the resonant circuit B via line 104. Phaseshifting circuitry is present in driving circuit C to shift the sensedcurrent signal by a predetermined phase angle. The phase angle φ may bean input to the system or the output of a voltage compensator, e.g.circuit 270 (FIG. 8). This reference voltage V_(ref) could also be usedas a dimming interface in dimmable lamp systems. The driving signals 106and 108 to the inverter are based upon this phase-shifted current signalto provide zero-voltage switching and to control inverter circuit A.

The voltage across the load is present on line 105. This signal is usedto provide voltage feedback and regulation. Voltage across the load iscompared with a reference voltage V_(ref) in driving circuit C as willbe described below.

As is known in the art, the normalized frequency Ω (i.e. Ω=ω_(s) /ω_(r),where ω_(r) is the resonant frequency and ω_(s) is the operatingfrequency) and the quality factor Q uniquely determine the second ordercircuit characteristics. Thus, when the quality factor Q is known, thepower source frequency Ω is traditionally used to control the circuitvoltage gain A, since A=f(Ω,Q). Likewise, the input phase angle φbetween the voltage and current is a function of the operatingfrequency, i.e. =g(Ω,Q). Thus, for a particular quality factor Q we canfind an inverse function for frequency, i.e. Ω=h(φ), in the region wherethe relation between φ=g(Ω,Q) andΩ is one-to-one. That is, there is aunique solution in which there is one operating frequency for aparticular phase angle. The function in this region is plotted in FIG.2(a). Substituting the function Ω(φ) into the known expression forcircuit voltage gain A(Ω) results in a relation A(φ), which is plottedin FIG. 2(b). Therefore, it is possible to regulate the voltage gain Aby using the input phase angle φ as a control variable.

As described above, there may be a region in which the relation betweenphase angle φ and frequency Ω is not one-to-one. Thus, for each chosenφ, there may be two or more corresponding frequencies. In order to usethe phase shift based control in this region, some measures need to betaken to eliminate either the higher frequency branch or the lowerfrequency branch. Properly designed filters could be used to serve thispurpose, as will be described hereinbelow.

In the implementation of the phase shift based control method, the inputphase angle φ is the driving variable and the source frequency is aderived variable along the curves of FIGS. 2(a) and 2(b). An open orclosed loop system may be used to implement the phase shift basedcontrol scheme.

The bridge type inverter output voltage V_(s) and the current flowingthrough the resonant circuit i_(L) are plotted in the time domain inFIG. 3. The phase angle between the voltage V_(s) and current i_(L) isφ. The phase angle is defined as φ=2π(t.sub.φ /T_(s)), where t.sub.φ issuch that i_(L) (t.sub.φ)=0, or the zero current crossing point. Bysensing the current signal i_(L), processing it with a driving circuitand driving the switches of the inverter, a feedback loop isimplemented.

With the circuit topology as in FIG. 7, the voltage gain and the phaseangle are plotted versus frequency, see FIGS. 4(a) and 4(b), for a lampload configuration. As described above, the inverter is favorablyoperated in the higher frequency branch of the voltage gain frequencycurves where the input phase angles are kept positive.

When the load changes, the steady state operating point shifts in orderto achieve a regulated output voltage. For a step change in the load,the steady state operating point shifts from one curve to another. Inorder to regulate the output voltage to a constant value, thecorresponding frequency has to be changed. In some cases, therelationship is strongly dependent on the load so that the voltagegain/frequency curves for different loads are widely spread, e.g. whenthe lamp resistance changes due to the dimming operation. Traditionalfrequency based control method has difficulty in reaching the properoperating points during fast load change. Moreover, the power switchesin the prior art system tend to lose zero-voltage switching during theprocess.

The voltage gain is plotted versus phase angle in FIG. 5 based on therelationships of voltage gain to frequency in FIG. 4(a) and phase angleto frequency in FIG. 4(b). When the load changes, the voltage gainversus frequency and the input phase angle versus frequency curves willchange accordingly. At a first load condition illustrated by curve LO1,the circuit operating point is at operating frequency f_(A), voltagegain V_(A) and phase angle φ_(A). The phase angle φ_(A) is positivewhich indicates the zero-voltage-switching condition for all bridgepower switches, i.e. Q1 and Q2. When the load changes abruptly, thecircuit characteristics change from curve LO1 to LO2 in FIGS. 4(a), 4(b)and 5. The traditional frequency sweeping based control method will movethe operating point to an intermediate point having the same initialoperating frequency f_(A), but new voltage gain V_(B') and new phaseangle φ_(B') before sweeping to the new steady state point at operatingfrequency f_(B), voltage gain V_(B) and phase angle φ_(B). At thisintermediate point, the phase angle φ_(B') is negative. Therefore, thezero-voltage switching condition is not maintained, which in turn maycause switch failure. By implementation of the phase angle shift controlscheme, the phase angle can be adjusted accordingly such that the outputvoltage is regulated, as illustrated in FIG. 5, Here, the operatingfrequency is adjusted indirectly. The phase angle change region remainsat the positive side. This implies that the inductive mode is alwaysmaintained, even during the transient process. More specifically, inorder to maintain a constant voltage gain, the values ε1 and ε2 of FIG.3 need to be adjusted as described below.

A proposed relation H for the driving circuit is shown in FIG. 6. Thetrigger points a and b in FIG. 6 correspond to the points a and b inFIG. 3. The variables ε1 and ε2 are selected as the hysteresis thresholdof FIG. 6, such that i_(L) (t=0)=ε1 and i_(L) (t=DT_(S))=ε2 in FIG. 3.The time duration from the zero current crossing point i_(L) =0 to thetrigger point a when i_(L) is decreasing is Δt₁, and the duration fromthe zero current crossing point when i_(L) is increasing to the triggerpoint b is Δt₂. Thus, the switching in the inverter occurs when thecurrent i_(L) crosses the values ε1 and ε2.

The quantities of ε1, ε2 and Δt₁, Δt₂ are related and are a function ofthe input phase angle φ. As ε1 or ε2 is decreased, the switchingthreshold is reached sooner. Consequently Δt₁ or Δt₂ decreases, and theswitching frequency increases. Likewise, the converse is typically truethat an increase in ε1 or ε2 results in a decrease in frequency.

In the power inverters, the load may change during the operation. If His fixed, a new stable point may be reached as long as some necessaryand sufficient conditions are met. The power inverter characteristics(oscillating frequency, gain, phase angle, etc.) change with the load.If such a change is not favorable to certain performance objectives, anecessary control scheme is needed. In this case, we can control theoperation by adjusting the hysteresis H in FIG. 6. The control variablescould be ε1 and ε2, which are used to effectively change the phase angleφ. The hysteresis is implemented in the preferred embodiment as acomparator circuit as illustrated in FIG. 10, although other methods arecontemplated as within the scope of the invention. It is likewisecontemplated that the hysteresis control may be implemented inconjunction with the phase-shift circuitry. For example, the sensedcurrent signal may be phase-shifted and subsequently the phase shiftedsignal is used to create a driving signal by use of a hysteresisfunction as above.

The Inverter Circuit

Turning now to FIG. 7, inverter circuit A includes a pair of switches Q1and Q2 which are arranged in a half-bridge configuration and convert theDC voltage from voltage source V to a high-frequency substantiallysquare wave AC output signal across the inverter outputs, under thecontrol of driver circuit C. Although the present embodiment isdescribed with respect to a half-bridge inverter having a pair ofswitches, the switching scheme described herein is applicable to a fullbridge topology having four switches. In such a configuration, theswitches would be controlled in pairs rather than individually

The switches Q1 and Q2 are preferably MOSFETs, although it iscontemplated that BJTs could also be used as the power switches. SwitchQ1 is effectively connected in parallel with diode D1, which could bethe body diode if MOSFETs are used. A snubber capacitor may also beconnected in parallel with switch Q1, which may include the junctioncapacitor incidental to switch Q1. Similarly, switch Q2 is connected inparallel with diode D2. Likewise, a snubber capacitor may also beconnected in parallel with switch Q2, which may include the junctioncapacitor incidental to switch Q2. The drain of switch Q1 is connectedvia the rail RL1 to the output from the voltage source V. The source ofthe switch Q1 is connected to the drain of switch Q2. The control gateof switch Q1 is connected via control line 108 to a respective gatecontroller terminal of driving circuit C via a parallel arrangement of aresistor R10 and a diode D10. The anode of diode D10 is connected to thecontrol gate of the switch Q1. The diode D10 provides rapid evacuationof charges from the control gate to enhance switching speed. The controlgate of switch Q2 is similarly connected to a gate controller terminalof driving circuit C through control line 106. A similar parallelarrangement exists for switch Q2 with resistor R12 and diode D12. Thisparticular arrangement has been implemented in conjunction with MOSFETswitches Q1 and Q2. A different gate drive would be implemented forbipolar transistor switches, which is within the scope of the presentinvention

Line 100 connects the midpoint I between the source of the switch Q1 andthe drain of the switch Q2 to one end of the blocking capacitor Cbk.Capacitor Cbk blocks DC components of the inverter output voltagegenerated at node I. Line 112 connects the midpoint I via resistor R14to driving circuit C, as will be described below Line 102 is connectedto the source of switch Q2. Line 100 and line 102 serve as the output ofinverter circuit A. The output of inverter A is a substantially squarewave voltage source.

Resonant Circuit

With continued reference to FIG. 7, the resonant circuit B shapes thevoltage and current waveforms to the load, such as a parallel loading ofan instant start fluorescent lamps L1, which is in series withballasting capacitor CL1.

The resonant circuit B includes resonant inductor Lr connected in serieswith the blocking capacitor Cbk via line 100. Resonant capacitor Cr isconnected in parallel with a primary winding of an isolation transformerT1.

Due to the high Q filtering action of the resonant circuit B on thesquare wave voltage output of the inverter circuit A, the current flowswith a substantially sinusoidal waveform. This current is sensed by thevoltage drop on line 102 across resistor R20, having a small resistance.The sensed current signal on line 104 serves as the input to drivingcircuit C as will be described below.

Inverter Controller Circuit

FIG. 8 illustrates, in block diagram form, the components of drivingcircuit C. Current sensing circuit 200 senses the voltage drop acrossR20. High-pass filter 204 is used to filter one of the frequencycomponents of the sensed current. The higher frequency componentcorresponding to the operating frequency is permitted to pass, but lowerfrequency signal components are suppressed from the current signal. As aresult, the inverter is forced to operate in the higher frequency branchof the voltage gain frequency curves, where the input phase angles arekept positive. Automatic gain control 206 is utilized to control theamplitude of the signal so as to avoid saturating circuit components.Phase shifting circuitry 208 provides a phase shift of φ to the inputsignal. The amount of phase shift φ applied by phase shifting circuitry208 is determined within a closed loop feedback scheme. The phase shiftcontrol voltage is the controlled output on line 105. The feedback pathconsists of feedback gain circuit 250. The reference voltage V_(ref) ispreselected and compared with the actual voltage in compensator 270,which provides an actuating signal on line 150. The signal on line 150is used to control the amount of phase angle shift to regulate theoutput voltage. Wave shaping circuitry 210 is triggered by thezero-crossing points of the phase-shifted current signal. Pulse widthmodulation 212 produces the switch pulses, and gate driver 216 controlsthe timing of the switches and corresponding duty cycles. Start-up logic214 transitions operation of the lamp from start-up to the steady stateoperating condition.

Turning now to FIG. 9, current sensing circuit 200, a differentialamplifier, senses the voltage drop across R20. R22, R24, R26, R28 andR30 are connected to op-amp 202, such as LT1122, to provide an outputvoltage signal on line 106 which is proportional to the current in theresonant circuit B. The signal is substantially sinusoidal at theoperating frequency, and normally has lower frequency components. Highpass filter 204 receives the signal of the sensed current. Capacitor C10and resistor R32 are selected to suppress this low frequency componentof the current signal. The present invention is applicable to resonantcircuits of higher order as well. Therefore, there may be a plurality oflow frequency components of the oscillating current signal which aresuppressed by filter 204.

Automatic gain control 206 is provided on the output from high passfilter 204, which prevents the sensed current signal from saturating thecomponents of the following circuitry. A pair of inverting op amps 220and 222 are used to scale the signal magnitude and provide a positivegain factor. The ratio of external resistors R34/R36 for op amp 220 andR38/R40 for op amp 222 are selected to scale the signal. Because theamplitude of the signal flowing through the circuit may fluctuate, anautomatic gain controller 224 is used to maintain the signal within agiven range. Automatic gain controller 224 is followed by op-amp 226.The gain is controlled by voltage controlled resistor, such as JFET T1,in parallel with impedance R42. The gate of JFET T1 is connected to thefeedback path consisting of impedance R44, diode D20, zener diode D22and an RC circuit consisting of resistance R46 and capacitance C12. Anadditional inverting amplifier 226 further scales the signal byselection of resistors R52 and R54.

Phase shifting of the sensed signal is performed by the phase-shiftingcircuitry 208, which includes the pair of all-pass filters 228 and 230in series. The all-pass filters incorporate op-amps such as LT1122. Asis known in the art, such filters provide a loop gain of 1 but provide apredetermined phase angle shift to the signal passing therethrough. Theamount of phase shift angle is determined by the control voltage on line150, which is the output of the voltage compensator 270.

The circuitry of the present invention allows the phase angle to be adriving variable. As described above, the application of a positivephase angle to the sensed current signal ensures zero-voltage switchingwhen the zero-crossing points of the phase shifted signal is used as acontrol signal for turning on the switches in the inverter circuit A.Ideally, the selection of any positive phase angle will assurezero-voltage switching. In practice, however; due to the output junctioncapacitance of the MOSFETs used for switches Q1 and Q2, there is aminimum phase angle requirement.

The phase angle may be implemented by an open loop or a closed loopbased scheme. In the open loop scheme (not shown), the amount of phaseshift may be preselected by selecting the amount of resistance R56 andR58 in parallel with the capacitors C14 and C16, respectively; to thenon-inverting input in all-pass filters 228 and 230, respectively. Suchresistors R56 and R58 may be at ground potential (not shown).Alternatively, potentiometers (not shown) or voltage controlled devices,such as JFETs T10 and T12, may be used in the closed loop based scheme.The present circuit is shown with two all-pass filters, although agreater or fewer number of filters is contemplated. The optimum phaseshift may be determined empirically from the circuit characteristics,such as the capacitance of the snubbing capacitors in parallel withswitches Q1 and Q2 if used in the circuit, or the current.

In order to regulate the output voltage, a closed loop scheme may beimplemented including voltage compensation circuitry, as illustrated inFIG. 9A. As described above, voltage regulation is possible by using thephase angle. With such a scheme, output voltage regulation may beperformed by sensing the output voltage across the load. In the presentembodiment, voltage across the load is present on line 105. The signalis rectified by diode D30 and scaling is applied by resistors R80 andR82 in series. Filtering of the signal is provided by the filter 252,including series resistors R84 and R86 in parallel with capacitor C40.The signal is amplified by noninverting amplifier 254, consisting ofop-amp 256, with the ratio of resistance R88/R90 chosen to provide theproper gain.

The signal on line 140 subsequently is acted upon by the voltagecompensator circuit 270. The reference voltage is applied to op-amp 272at the noninverting terminal. Reference voltage V6 may be applieddirectly. Preferably, a voltage divider including R92 and potentiometerR94 may be used to provide a variable reference voltage. The op-amp 272may be configured as a PI regulator, with the feedback path having aparallel combination of capacitor C42 with resistor R96 and capacitorC44 in series. The output of the PI regulator encounters resistor R98and the parallel combination of diode D40, zener diode D42 and capacitorC46, which limit the dynamics of the signal in order to avoid saturatingthe components of the circuitry, namely JFETs T10 and T12 (FIG. 9), aswill be described below.

With continued reference to FIG. 9, the output signal of compensator 270is present on line 150. JFETs T10 and T12 are in series with resistorsR58 and R56 respectively. Line 150 is connected to the gates of T10 andT12 and thereby controls the resistance at the noninverting terminals ofthe op-amps 228 and 230. Thus the phase angle shift φ may be preciselycontrolled.

Since the output voltage is a function of the operating frequency or thephase angle, output voltage regulation may be achieved by adjusting thephase angle in order to produce the desired output voltage. This schememay find application in light dimming ballasts.

Following the phase shifting of the signal, the wave shaping circuitry210 provides timing for the initiation of the switching as will bedescribed below. A filter including capacitor C18 shapes the signal, andbuffer 231 is a voltage follower interposed to provide a low outputimpedance and isolation from the input. The output of buffer 231 is online 119. Noninverting amplifier 232 (FIG. 10) includes op-amp 234, andexternal resistors R60 and R62 are selected to scale the voltage. Filter236 includes resistor R64 and capacitor C20.

With continued reference to FIG. 10, the switching signal is generatedby a timing circuit, such as a standard 555 timing IC 300, an exemplarycircuit being produced by Motorola, for example, the specifications ofwhich are incorporated by reference herein and illustrated in FIG. 10A.Timer 300 is configured in the monostable mode wherein the drivingsignal is triggered to the "on" state by the zero-crossing of thephase-shifted current signal. Comparator 310 converts the substantiallysinusoidal analog voltage signal from filter 236 to a square wavesequence which corresponds to the zero-crossing point of the shiftedcurrent signal. The comparator converts this zero-crossing point to setthe leading edge of the switching pulse.

Applied voltage V1, in combination with resistors R66 and R68 and diodeD30, provide a proper scaling to the signal applied to the inverter I1.High pass filter 238, including capacitor C22 and resistor R70 and diodeD32 in parallel, and inverter I2 further shape the signal on line 120,which is connected to the trigger input (pin 2) of the timer 300. Thethreshold terminal (pin 7) and the discharge terminal (pin 6) areconnected together in this mode. The control voltage at pin 5 is presetto a constant value by a voltage divider with resistors R72 and R74. Theoutput signal is present on line 122 from output terminal (pin 3).Output signal on line 122 from timing circuit 300 and output on line 123from start-up circuit 320 pass through OR gate 214 which acts as anelectronic switch.

As illustrated in FIGS. 11-12, start-up circuit 320 provides thefunction of initial frequency sweeping for lamp ignition. Start-upcircuit 320 includes voltage controlled oscillator (VCO) 312, singlepulse flip-flop 314 in FIG. 11 and a DC voltage sensing and scalingcircuit 316 (FIG. 12). VCO 312 is preferably a voltage-to-frequencyconverter AD654 produced by Analog Devices of Norwood, Mass., thespecifications of which are incorporated by reference herein, althoughany suitable VCO is contemplated for use in this circuit. The start-upprocess is described as follows. In FIG. 11, before the DC bus voltage Vis fully built up, the voltage on line 130 is low and the output pin Qof the flip flop 314 is high. VCO 312 starts to operate at highfrequency region f_(start) which leads to a high frequency square wavesignal on line 123. The output voltage of the resonant circuit B is lowdue to the high frequency operation and lower DC bus voltage V.

When the DC bus voltage is increased to a preset threshold, the outputvoltage of comparator 318 in FIG. 12 is set to low. This in turntriggers the flip flop 320 and sets the voltage to high at line 130.Effectively, in FIG. 11, the voltage input at control voltage terminal(pin 4) of VCO 312 is reduced gradually. Correspondingly, the outputfrequency produced by VCO 312 is reduced. This process completes thefrequency sweeping from high frequency f_(start) to lower operatingfrequency f_(op). Correspondingly, the resonant circuit output voltageto the lamp L1 is gradually increased until the lamp is ignited. Afterthe lamp is operating, the processed current signal in line 122 triggersthe flip-flop 314 and the output signal on line 123 is blocked byoperation of the OR gate 322. Subsequently, the power supply in FIG. 8operates at a self-oscillating frequency at a predetermined phase angleφ.

Turning now to FIG. 13, the signal on line 124 is inverted at logicalinverter I6. The signal subsequently passes to the "in" terminal of gatedriver 330, such as a half-bridge driver integrated circuit IR2111produced by International Rectifier, El Segundo, Calif., thespecifications of which are incorporated by reference herein andillustrated at FIG. 13A. With continued reference to FIG. 13, gatedriver 330 includes logic to convert a single square wave input signal,such as that on line 124, into two separate driving signals on lines 108and 106 which control switches Q1 and Q2, respectively. Gate driver 330may be configured to produce signals 108 and 106 such that a blankingtime exists in which neither Q1 nor Q2 conduct.

The waveforms produced by the above-described circuit are represented inthe time plot illustrated in FIG. 14. The current flowing through theresonant circuit B is represented on the plot by signal 500. Withindriving circuit C, this signal corresponds to the voltage signal on line114, which is proportional to the current and has been filtered of lowerfrequency components by high-pass filter 204 (See, FIGS. 8-9). Asdescribed above, the signal on line 114 passes through automatic gaincontrol 206 and phase-shifting circuitry 208. The phase shifted signal502 (FIG. 14) corresponds to the output of phase-shifting circuitry 208on line 118 (See, FIG. 9). The phase-shifted signal 502 is phase-shiftedby an amount 504 corresponding to φ as described above. Thezero-crossing point of signal 502 is used as the reference to controlswitching. Thus, at the zero-crossing point 508, switch Q1 is turnedoff, as indicated by line 510 (solid line) and switch Q2 is turned on,as indicated by line 512 (dot-dash line). The figure illustrates that Q2is turned on at the same time that Q1 is turned off. A blanking time 514is ordinarily provided by driver 304 such that Q1 and Q2 are both offfor a brief period of time. When the system load changes, the voltagecompensator 270 in FIG. 9(a) generates an appropriate control signal online 150 of FIG. 9(a), which is effectively connected to the gates ofJFETS T10 and T12 in FIG. 9, As a result, the phase angle φ is adjustedaccordingly such that the output voltage is regulated.

An alternative embodiment is shown at FIG. 15. A controllable hysteresisthreshold may be implemented as described above with respect to FIGS. 3and 6. Comparator 310 converts the substantially sinusoidal analogvoltage signal from op-amp 226 to a square wave sequence whichcorresponds to the crossing points of the current signal with thethreshold settings ε1 and ε2. The logic selection unit 320 converts thebasic unipolar DC signal in line 150 into a bipolar DC signal at line151, which serves as the threshold value ε1 and ε2 as in FIG. 3. Thecomparator detects these crossing points to set the leading edge of theswitching pulse. In such an implementation, the converted voltagesensing on line 150 of FIG. 9A and the filtered current signal on line116 of FIG. 9 are inputs to the comparator. In particular, the voltagesignal 150 via the logic selection unit 320 would be fed into thenon-inverting input of the comparator 310 and the current signal 116would be fed into the inverting input of the comparator 310. It is alsocontemplated that the phase-shifting circuitry be applied as shown inFIGS. 9 and 9A to the current signal and a hysteresis control be used inconjunction therewith to adjust the driving signal and consequently theswitching times. It is further contemplated that a Schmitt trigger orother voltage level detector may be used in the circuit as analternative to the comparator 310.

FIGS. 16 and 17 illustrate second and third alternative embodiments ofthe subject invention. In these embodiments, the hysteresis threshold,e.g. ε1 and ε2, is fixed, rather than controllable. Thus, a constantvoltage V10 and V12, through a logic selection unit 320, is applied tothe non-inverting input of comparator 310 in the embodiments of FIG. 16and FIG. 17, respectively. It is also contemplated that voltages V10 andV12 may be variable. The circuits are substantially as described abovewith respect to the previous embodiments, with the differences notedhereinbelow. In the second embodiment of FIG. 16, the output voltagesignal of the voltage compensator 270 on line 150 (FIG. 9A) and thesensed current signal on line 115 (FIG. 9) are multiplied in multiplyingIC 360, and the product of the two signals is directed to the invertinginput of comparator 310. By doing so, the relative magnitude between thesensed current signal and the threshold ε1 and ε2 is changed withreference to FIG. 3, In this case, the thresholds ε1 and ε2 are fixedwhile the sensed current signal is multiplicatively scaled up or downcontrolled by the output signal on line 150 of the voltage compensator270. Consequently, the switching time Δt₁ and Δt₂ are adjusted in FIG.3. In return, the phase angle φ is controlled.

In the third embodiment of FIG. 17, the control voltage signal on line150 (FIG. 9A) and the sensed current signal on line 115 (FIG. 9) areadded via the combination of multiplying IC 370 and op-amp 270. By doingso, the relative magnitude between the sensed current signal and thethresholds ε1 and ε2 is changed with reference to FIG. 3. In this case,the thresholds ε1 and ε2 are fixed while the sensed current signal ismultiplicatively and additively scaled up and down controlled by theoutput signal on line 150 of the voltage compensator 270. This meansthat the switching times Δt₁ and Δt₂ are adjusted as in FIG. 3. Inreturn, the phase angle φ is controlled.

In another alternative embodiment, it is contemplated that output power,rather than voltage alone, may be controlled. For example, lamp powermay be regulated for applications such as dimming. The phase angle shiftcontrol method described above for voltage regulation may be used forpower regulation with the following modifications. The lamp power signalis sensed and fed back to the control circuits. In a preferredembodiment, a voltage sensing line is positioned at the anode of thelamp L1 to sense lamp voltage. By adding a small resistor in series withthe lamp and sensing the voltage across the inserted resistor, lampcurrent information may be obtained. By further multiplying the sensedlamp voltage and current signal using a standard analog multiplier IC, afeedback signal is produced which is proportional to the lamp power andis fed back to the control circuit compensator. A reference lamp outputpower is selected. By comparing the feedback signal with the referencepower, a compensator provides a phase shift command to the phaseshifting circuitry. In such a way, the lamp output power is regulated.

While there has been shown to be what are presently considered to be thepreferred embodiments of the invention, it will be apparent to those ofordinary skill in the art that various modifications can be made withoutdeparting from the scope of the invention as defined by the appendedclaims. Accordingly, the disclosure is illustrative only and notlimiting.

What is claimed is:
 1. A power supply, which comprises:a) a resonantcircuit having an output voltage and a current oscillating therethrough;b) a voltage-fed half-bridge inverter producing a supply voltage at anoutput coupled to the resonant circuit, the inverter being responsive toa driving signal; and c) a driving circuit producing the driving signal,the driving circuit comprising:i) voltage compensation circuitryregulating said output voltage with respect to a reference voltage andcommanding a phase shift angle; and ii) phase-shifting circuitryapplying a phase-shift to a sensed current signal, the amount of phaseshift being commanded by the voltage compensation circuitry.
 2. A powersupply as claimed in claim 1, wherein the driving circuit furthercomprises circuitry for comparing the phase-shifted sensed currentsignal with at least one reference.
 3. A power supply as claimed inclaim 2, wherein the one reference is a function of the phase-shiftangle.
 4. A power-supply as claimed in claim 2, wherein the circuitryfor comparing the phase-shifted sensed current signal with the at leastone reference comprises a hysteresis relationship.
 5. A power supply asclaimed in claim, 2, wherein the driving circuit further comprises meansfor scaling the sensed current signal with respect to a voltage signalof the voltage compensator.
 6. A power supply as claimed in claim 2,wherein the driving circuit further comprises means for adding thesensed current signal with an output signal of the voltage compensator.7. A power supply, which comprises:a) a resonant circuit having anoutput voltage, an output power, and a current oscillating therethrough;b) a voltage-fed inverter producing a source voltage at an outputcoupled to the resonant circuit, the inverter being responsive to adriving signal; c) means coupled to the resonant circuit for deriving asensed current signal determined by the current flowing in the resonantcircuit, and d) a driving circuit producing the driving signal, thedriving circuit comprising:i) power compensation circuitry regulatingsaid output power with respect to a reference power signal andcommanding a phase shift angle; and ii) phase-shifting means applying aphase-shift to the sensed current signal with the amount of phase shiftcommanded by the power compensation circuitry.
 8. A method for switchinga power supply, which comprises:providing a resonant circuit having acurrent oscillating therethrough and an output voltage; providing aninverter responsive to a driving signal and producing a voltage at anoutput coupled to the resonant circuit; detecting the output voltage andcomparing the output voltage with a reference voltage; commanding aphase-angle shift to reduce the difference between the output voltageand the reference voltage; applying the phase-angle shift to a signalrepresenting the current oscillating through the resonant circuit andproducing a phase-shifted current signal; and driving the inverter basedupon the phase-shifted current signal.
 9. A method for switching a powersupply as claimed in claim 8, wherein the step of driving the inverterfurther comprises:comparing the phase-shifted current signal with ahysteresis threshold.
 10. A method of switching a power supply asclaimed in claim 8 which further comprises actuating a switching devicein the inverter when the phase-shifted current signal crosses a zerovalue.
 11. A power supply as claimed in claim 1 wherein thephase-shifting circuitry applies a positive phase angle shift to thesensed current signal such that the inverter output leads the sensedcurrent signal with a predetermined positive phase angle.
 12. A powersupply as claimed in claim 1 wherein the driving signal produced by thedriving circuit actuates a transistor switch in the inverter when thephase-shifted sensed current signal crosses zero.
 13. A power supply asclaimed in claim 1 wherein the phase-shifting circuitry includes anall-pass filter and the voltage compensation circuitry is responsive toa feedback signal determined by the output voltage of the resonantcircuit.
 14. A power supply as claimed in claim 1 wherein the invertercomprises first and second transistor switches connected in seriescircuit across a pair of terminals which supply a DC voltages, andtheresonant circuit comprises a first inductor and a capacitor connected inseries to a circuit node intermediate the first and second transistorswitches and a second inductor connected in parallel with the capacitorand adapted to be coupled to a load.
 15. A power supply as claimed inclaim 1 wherein the sensed current signal is determined by andrepresents the current oscillating in the resonant circuit.